Switching Voltage Transient Protection Schemes For High-Current Igbt Modules, elektronika, elektronika, ...
[ Pobierz całość w formacie PDF ]
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 33, NO. 6, NOVEMBER/DECEMBER 1997
1601
Switching Voltage Transient Protection Schemes
for High-Current IGBT Modules
Rahul S. Chokhawala,
Member, IEEE
, and Saed Sobhani
Abstract—
The emergence of high-current and faster switching
insulated gate bipolar transistor (IGBT) modules has made it
imperative for designers to look at ways of protecting these
devices against detrimental switching voltage transients that are a
common side effect of these efcient transistors. This paper will
discuss protection criteria for both normal switching operation
and short-circuit operation and will cover in detail some of the
protection schemes that were designed to address these problems.
Index Terms—
Insulated gate bipolar transistor (IGBT), protec-
tions circuits, resistor–capacitor diode (RCD), short-circuit safe
operating area (SCSOA), short circuit, switching safe operating
area (SSOA), switching transients, voltage clamps.
I. I
NTRODUCTION
W
HEN a power device is abruptly turned off, trapped
energy in the circuit stray inductance is dissipated in
the switching device, causing a voltage overshoot across the
device. The magnitude of this transient voltage is proportional
to the amount of stray inductance and the rate of fall of turnoff
current. Large insulated gate bipolar transistor (IGBT) modules
switch high magnitude of currents in a short duration of time,
giving rise to potentially destructive voltage transients. These
higher current modules normally consist of several IGBT chips
in parallel. Each individual chip switches its share of the load
current at a that is determined by the gate drive circuit.
The total current and seen by the external power circuit
is the sum of currents and ’s through each IGBT chip.
The situation is at its worst when a short-circuit current is
rapidly turned off to protect the IGBT. The ’s produced
could easily be a few thousand A/ s. If proper attention is not
paid to minimize resulting switching voltage transients, any
attempt to save IGBT’s, by shutting them down under fault
conditions, may destroy the device.
This paper discusses various protection schemes. A tran-
sient voltage protection scheme optimized to protect IGBT’s
during normal switching operation may not protect the IGBT’s
under fault current shut-off process. Separate schemes would
normally be required to achieve both goals.
Fig. 1. Rated SOA curve and operating loci with and without switching
voltage transient protection circuit.
It is determined that the snubbers and clamps offer op-
timized protection against voltage transients during normal
switching operation. Operation of a resistor–capacitor diode
(RCD) clamp circuit is described in detail. As illustrated in
Fig. 1, protection circuits allow faster yet safer operation by
containing operating loci within the boundaries of the rated
safe operating area (SOA).
Fault current shut-off transients are more effectively pro-
tected by considerably slowing the rate of fall of fault current.
Two novel protection schemes are introduced which protect
IGBT’s from potentially destructive voltage transients by
slowing the rate of fall of fault current, only under fault
conditions. Circuit operations are analyzed and the test results
are illustrated. Usefulness of an active clamp is also discussed
in this section.
II. V
OLTAGE
T
RANSIENTS
D
URING
N
ORMAL
S
WITCHING
O
PERATION
As mentioned earlier, the magnitude of transient voltage
depends on the trapped energy in the circuit stray inductance,
also called “dc loop” inductance As a preventive measure,
steps should be taken to improve the circuit layout. Usage of
copper plates separated by a thin sheet of insulating material,
tightening the “dc loop,” and choosing source capacitance
with inherently low self inductance are ways to lower stray
inductances [1]. Decoupling capacitors, connected across the
module terminals, can also be used to achieve this goal. High-
Paper IPCSD 97–32, presented at the 1994 IEEE Applied Power Electronics
Conference and Exposition, Orlando, FL, February 13–17, and approved for
publication in the IEEE T
RANSACTIONS ON
I
NDUSTRY
A
PPLICATIONS
by the
Power Electronics Devices and Components Committee of the IEEE Industry
Applications Society. Manuscript released for publication May 9, 1997.
R. S. Chokhawala is with ABB Semiconductors AG, CH-5600 Lenzburg,
Switzerland.
S. Sobhani is with International Rectier Corporation, El Segundo, CA
90245 USA.
Publisher Item Identier S 0093-9994(97)08440-5.
0093–9994/97$10.00
ã
1997 IEEE
1602
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 33, NO. 6, NOVEMBER/DECEMBER 1997
Fig. 2. RCD voltage clamp.
Fig. 4. Turnoff waveforms with and without RCD snubber. Tested at: 400 V,
100 A, 25
C;
L
s
=
100 nH;
V
G
=R
G(o)
=
-8 V/33
;
C
sn
=
0.22
F;
R
sn
=
12
;
V
CE
: 100 V/div;
I
C
=I
sn
: 20 A/div;
E
o
: 2 mJ/div.
also for reducing IGBT turnoff losses. During IGBT turn-on,
the snubber capacitor is fully discharged, and during turnoff,
it is charged. This circuit, unlike the circuit in Fig. 2 which
essentially acts as a clamp, reduces the rate of rise of voltage
across the IGBT at turnoff, imposing a softer switching and,
therefore, reducing losses in the IGBT. The losses in the
snubber, however, are substantially increased and are equal to
where is the voltage across snubber capacitor
at the end of turnoff process and is equal to the dc bus plus
an allowable overshoot voltage.
Due to the dual purpose that the circuit in Fig. 3 serves, the
tradeoffs involved are complex. Since this paper concentrates
only on the switching voltage transient protection, discussion
will be focused on the circuit in Fig. 2. The effects of this
snubber on turnoff and turn-on will be discussed separately in
the following.
Fig. 3. RCD charge/discharge snubber.
frequency polypropylene capacitors designed for low internal
lead inductance are found to be effective. Care should be
taken in the selection of the decoupling capacitor value to
avoid oscillations in the dc loop which otherwise may result
in excessive heating in the high-frequency capacitors. For
modules rated up to 100 A or so, decoupling capacitors may
provide optimal protection against voltage transients during
normal switching.
One other way to prevent high-voltage transients from
occurring is to slow down the switching process, by choosing a
greater value of gate resistor. While this is an attractive method
for the fault current turnoff protection, it is not practical for
protection against voltage transients during normal switching
operation, as the efciency of device operation is adversely
affected.
In the following discussion, more efcient ways of protect-
ing devices will be presented.
A. Turnoff
The RCD clamp of Fig. 2 acts as a voltage clamp. During
IGBT conduction period, the snubber capacitors are charged
to the bus voltage. As the IGBT is turned off, the voltage
across it, , rises rapidly. The circuit “dc loop” stray
inductance may cause to rise above the bus voltage.
As this occurs, the snubber diode is forward biased, and
the snubber is activated. The energy trapped in the stray
inductance now is diverted to the snubber capacitor, which
absorbs this incremental energy without substantial rise in its
voltage. The waveforms shown in Fig. 4 clearly illustrate the
turnoff behavior with and without the RCD clamp. The voltage
overshoot has been substantially reduced from 210 to only 50
V. Initially, a small stray inductance in the snubber circuit
causes to peak slightly above
Fig. 5 displays the waveforms generated for two different
stray inductances (100 and 340 nH). As illustrated in the
III. RCD S
NUBBER AND
C
LAMP
C
IRCUITS
Figs. 2 and 3 are two principal examples of RCD snubbers
for high-current IGBT applications. While both circuits are
employed to reduce transient voltages across switching de-
vices, the charge/discharge snubber circuit in Fig. 3 is targeted
CHOKHAWALA AND SOBHANI: SWITCHING VOLTAGE TRANSIENT PROTECTION SCHEMES
1603
Fig. 6. Turn-on waveforms for an IGBT with no RCD snubber protection.
Tested at: 400 V, 100 A, 25
C;
L
s
=
240 nH;
V
G
=R
G(on)
=
15 V/5.1
;
V
CE
: 100 V/div;
I
C
: 20 A/div;
V
diode
: 100 V/div.
Fig. 5. Turnoff waveforms with RCD snubber for two different
L
s
values
(100 nH, 340 nH). Tested at: 400 V, 100 A, 25
C;
V
G
=R
G(o)
=
-8 V/33
;
C
sn
=
0.22
F;
R
sn
=
12
;
V
CE
: 100 V/div;
I
C
: 20 A/div;
E
o
:
2 mJ/div.
gure, the initial peak—which is dependent on the
stray inductance within the snubber circuitry—is the same for
the two cases. The nal voltage peak for the higher
inductance does reach a higher value as expected, since there
is more trapped energy diverted to the same snubber
capacitor. This value, however, is well within the voltage
rating of the device and only marginally inuences the losses
in the IGBT, since it occurs when the current has reached to a
smaller value. The magnitude can be calculated from the
formulas given in the following section.
B. Turn-On
Fig. 6 displays the turn-on waveforms for an unprotected
IGBT with a gate resistor of 5.1 . The rapid rise in
the IGBT current (1200 A/ s) combined with the circuit stray
inductance (300 nH) caused the freewheel diode (FWD) to go
through severe reverse-recovery process. As seen in the gure,
the FWD recovery voltage (
Fig. 7. Turn-on waveforms for an IGBT with no RCD snubber protection.
Tested at: 400 V, 100 A, 25
C;
L
s
=
240 nH;
V
G
=R
G(on)
=
15 V/33
;
V
CE
: 100 V/div;
I
C
: 20 A/div;
V
diode
: 100 V/div;
E
on
: 1 mJ/div.
630 V) actually exceeded the
rated voltage of the module.
In order to bring this voltage down to a safe value, the turn-
on was lowered by using a higher The results are
shown in Fig. 7. The increase in however, had profound
effect in increasing the switching losses, as expected [2], [3].
The RCD clamp shown in Fig. 2 is also effective in reducing
turn-on voltage transients. As the IGBT current rises, the
voltage loss causes the voltage across the positive
and negative terminals of the module, to drop by the same
amount (i.e., to The snubber capacitors
that were fully charged to now nd a discharge path
through the forward-biased FWD (note that the FWD is on,
freewheeling the load current), the IGBT, and the snubber
resistors. Fig. 8 shows the equivalent circuit during turn-on.
The snubber diodes are reverse biased and, therefore, not
shown. The current paths are shown in the gure. This snubber
discharge current partially provides for the reverse-
recovery charge of the FWD, thus, the total current seen by
is modied. This has a favorable effect on the magnitude
of the reverse-recovery voltage transient.
The waveforms shown in Fig. 9 illustrate the snubber oper-
ation. Notice the complete elimination of the voltage transient
and, also, reduction in the oscillations following turn-on.
Another interesting fact is that this waveform was generated
with of 0.5 , which reduced the energy losses from 2.41
mJ in Fig. 7 to 1.25 mJ, a savings of almost 50%. Therefore,
this snubber not only clamps the turn-on voltage transient, but
also enables the user to choose a value of
that produces
minimal turn-on losses.
Fig. 10 shows the effect of changing the snubber resistor
on
turn-on
waveforms.
Lower
’s
provide
for
better snubbing action.
1604
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 33, NO. 6, NOVEMBER/DECEMBER 1997
Fig. 8. Equivalent
circuit
under
turn-on
conditions.
Low-side
IGBT
is
Fig. 10. Effect of changing
R
sn
values. Tested at: 400 V, 100 A, 25
C;
L
s
=
240 nH;
V
G
=R
G(on)
=
15 V/33
;
C
sn
=
0.22
F;
R
sn
=
12
,
33
;
V
diode
: 100 V/div;
I
total
: 20 A/div.
switched on.
Losses in the snubber resistor are
(3)
The snubber diode should be of fast and soft recovery
type to avoid severe oscillations following at turnoff. The
resistor should be of noninductive type to avoid oscillations
at turn-on.
IV. V
OLTAGE
T
RANSIENTS
D
URING
F
AULT
C
URRENT
T
URNOFF
The short-circuit current generated during fault conditions
can be up to ve to ten times the rated current. Shutting off
such high currents too quickly can produce extremely high
’s that are potentially detrimental to the IGBT’s [4].
The RCD clamp circuit discussed in the preceding section is
not as practical when it comes to protecting transient voltages
generated during short-circuit conditions. As seen from the
above expressions, the required snubber capacitor value is
proportional to the square of the device current. This means
that the capacitor required will have to be 25–100 times
larger than in normal switching operation. High-capacity high-
voltage snubber capacitors are large and expensive (as for
snubber capacitors required for GTO thyristors), making the
RCD scheme unattractive for high-current IGBT modules.
Also, voltage clamps connected externally to the modules
do
Fig. 9. Turn-on waveforms with RCD snubber. Tested at: 400 V, 100 A, 25
C;
L
s
=
240 nH;
V
G
=R
G(on)
=
15 V/0.5
;
C
sn
=
0.22
F;
R
sn
=
12
;
V
CE
=V
diode
: 100 V/div;
I
C
=I
sn
=I
total
: 20 A/div;
E
on
: 0.5 mJ/div;
V
ab
: 100 V/div.
The value for the snubber components can be approximated
from the expressions given below, based on circuit stray
inductance
switching frequency
maximum switching
current
turn-on current rise time
dc rail voltage
,
and allowable peak voltage
(see the Appendix).
not
address
the
problem of internal
inductive
voltage
The snubber capacitor is
spike.
In the following, we will discuss more practical methods.
This involves slowing down the turnoff of the IGBT’s under
fault conditions.
The IGBT fault current rate can be reduced by slowing the
turnoff gate voltage signal. The simplest way of achieving
this is to increase the gate resistor, but this is inefcient,
(1)
The snubber resistor is
(2)
CHOKHAWALA AND SOBHANI: SWITCHING VOLTAGE TRANSIENT PROTECTION SCHEMES
1605
Fig. 12. Short-circuit waveforms with resistive protection scheme for “fault
under load” condition. Tested at: 280 V, 25
C;
L
s
=
240 nH;
V
G
=R
G1
=
-8 V/33
;
V
CE
: 100 V/div;
V
GE
=
5 V/div;
I
sc
: 200 A/div.
Fig. 11. Circuit for the resistive method.
since the tradeoff is increased switching losses during normal
conduction.
In order to address the above problem, two novel circuits
are introduced that, through electronic gate control, effectively
decrease the rate of fall only when a fault current is
sensed, thereby avoiding any losses during normal switching
operation. The rst of these circuits utilizes a resistive method,
and the other uses a capacitive method. In the resistive method,
a considerably higher value of gate resistor is switched in,
in series with the IGBT gate. In the capacitive method, a
considerably higher value of external capacitor is switched
in, in parallel with the IGBT gate input capacitance.
Fig. 13. Short-circuit waveforms with no protection scheme for “fault under
load” condition. Tested at: 280 V, 25
C;
L
s
=
240 nH;
V
G
=R
G1
=
-8
V/33
;
V
CE
: 100 V/div;
V
GE
=
5 V/div;
I
sc
: 200 A/div.
V. R
ESISTIVE
M
ETHOD
The circuit in Fig. 11 is composed of de-sat sense diode
and a p-channel MOSFET to switch in higher value of resistor
upon occurrence of a fault.
Initially, when the IGBT is in the off state, the P-MOSFET
is turned off. During normal turn-on, a step rise in voltage
is applied to the IGBT gate through and the inherent
body diode of the P-MOSFET. As after normal turn-
on delay period, drops to its low on-state level, diode
is forward biased and input capacitance of the P-MOSFET
starts to charge up. During normal conduction, therefore, the
P-MOSFET remains gated on.
During normal turnoff operation, the gate drive output
voltage is switched to its low state. The P-MOSFET gate
capacitance begins to discharge. The values of and
are adjusted such that the MOSFET is kept on, at least until
the IGBT turnoff is completed (for example, 1
as the discharge is now forced to take place through
The fault current fall rate is decreased accordingly. Note that,
if the IGBT is turned off while the MOSFET is still on,
the circuit will not be effective, since is bypassed. The
above consideration places an upper limit on the discharge
time constant of the MOSFET (to, for example, 5 s).
Figs. 12 and 13 display short-circuit switching waveforms
with and without the protection circuit. The initial current
through the IGBT is 40 A. Upon occurrence of the fault, the
current shoots up to 800 A initially, but settles down to 600 A
once the Miller effect on the gate voltage is diminished (see
gate waveforms). The MOSFET discharge time constant was
adjusted to be 2.5 s Fault current
was turned off after 6 s. As seen from the aforementioned
gures, the IGBT gate discharge rate was considerably slowed
by the addition of (200 ), thereby reducing the voltage
overshoot from 270 down to 60 V.
The Miller effect can be ltered out by bypassing
with a diode. The IGBT gate voltage is now clamped to
the gate drive output voltage. Fig. 14 displays the resulting
waveforms. Compare the results to Fig. 12 (same protection
circuit without the bypass diode). The initial surge of
current is eliminated. The slight increase in the turnoff voltage
s). Therefore,
the IGBT turnoff losses are not affected.
A. Fault Under Load
When a fault occurs during normal conduction, diode
goes into blocking mode and the P-MOSFET input capacitance
starts to discharge through resistors and The MOSFET
is turned off as its gate voltage drops below the threshold
value. Thereafter, the
rate of fall is reduced signicantly
[ Pobierz całość w formacie PDF ]